diode capacitance changes, the feedback capacitor must
change to maintain a stable and flat frequency response.
Using Equation 1, CF is adjusted to give the Butterworth
frequency responses presented in Figure 4.
+5V
Power-supply
decoupling not shown.
RF
167Ω
0.01µF
VO = –
VI
OPA846
PHOTODIODE TRANSIMPEDANCE
RG
FREQUENCY RESPONSE
83
20 log(10kΩ)
RF = 10kΩ
–5V
CF Adjusted
80
77
74
71
68
65
62
CD = 10pF
RG
250Ω
RF
500Ω
VI
CS
27pF
CF
2.9pF
0Ω
Source
0.01µF
10kΩ
OPA846
VO = ID RF
CD = 100pF
CD = 50pF
RF
10kΩ
λ
ID
CF
CD
CD = 20pF
–VB
FIGURE 5. Broadband, Low-Gain, Inverting Amplifier.
1
10
Frequency (MHz)
100
Physically, this ZO (11.6MHz for these values) is set by:
1
FIGURE 4. Transimpedance Bandwidth versus CD.
2πRF C + C
(
)
F
S
LOW-GAIN COMPENSATION FOR IMPROVED SFDR
and is the frequency at which the rising portion of the noise
gain would intersect the unity gain if projected back to a 0dB
gain. The actual zero in the noise gain occurs at NG1 • ZO,
and the pole in the noise gain occurs at NG2 • ZO. Since GBP
is expressed in Hz, multiply ZO by 2π, and use this to get CF
by solving:
Where a low gain is desired, and inverting operation is
acceptable, a new external compensation technique may be
used to retain the full slew rate and noise benefits of the
OPA846, while giving increased loop gain and the associ-
ated improvement in distortion offered by the decompen-
sated architecture. This technique shapes the loop gain for
good stability, while giving an easily controlled 2nd-order
low-pass frequency response. Considering only the noise
gain (noninverting signal gain) for the circuit of Figure 5, the
low-frequency noise gain (NG1) is set by the resistor ratios,
while the high-frequency noise gain (NG2) is set by the
capacitor ratios. The capacitor values set both the transition
frequencies and the high-frequency noise gain. If this noise
gain (determined by NG2 = 1 + CS/CF) is set to a value
greater than the recommended minimum stable gain for the
op amp and the noise gain pole (set by 1/RFCF) is placed
correctly, a very well controlled, 2nd-order, low-pass fre-
quency response results.
1
CF =
= 2.86pF
(
)
(5)
2πRFZONG2
Finally, since CS and CF set the high-frequency noise gain,
determine CS by using NG2 = 10.5:
CS = NG − 1C , which gives C = 24.9pF
(6)
(
)
2
F
S
The resulting closed-loop bandwidth is approximately equal to:
(7)
f−3dB
ZO • GBP
For the values of Figure 5, f–3dB is approximately 142MHz.
This is less than that predicted by dividing the GBP product by
NG1. The compensation network controls the bandwidth to a
lower value, while providing the full slew rate at the output and
an exceptional distortion performance due to increased loop
gain at frequencies below NG1 • ZO. The capacitor values
shown in Figure 5 are calculated for NG1 = 3 and NG2 = 10.5
with no adjustment for parasitic components.
To choose the values for both CS and CF, two parameters and
only three equations need to be solved. The first parameter is
the target high-frequency noise gain (NG2), which should be
greater than the minimum stable gain for the OPA846. Here,
a target NG2 of 10.5 is used. The second parameter is the
desired low-frequency signal gain –(RF/RG), which also sets
the low-frequency noise gain NG1 (= 1 + RF/RG). To simplify
this discussion, target a maximally flat 2nd-order, low-pass
Butterworth frequency response (Q = 0.707). The signal gain
of –2 shown in Figure 5 sets the low-frequency noise gain to
NG1 = 1 + RF/RG (= 3 in this example). Then, using only these
two gains and the GBP for the OPA846 (1750MHz), the key
frequency in the compensation can be determined as:
See Figure 6 for the measured frequency response for the
circuit of Figure 5. This shows the expected gain of –2 (6dB)
with exceptional flatness through 70MHz and a –3dB band-
width of 170MHz. Repeating the swept frequency distortion
measurement for a 2VPP output into a 200Ω load and
comparing to the gain of +10 data shown in the Typical
Characteristic curves illustrates the improved distortion for
this low-gain compensation circuit.
Figure 7 compares the distortion at a gain of +10 for the
circuit of Figure 1 to the distortion at a gain of –2 for the circuit
of Figure 5.
GBP
NG21
NG1
NG2
NG1
NG2
ZO
=
1−
− 1− 2
(4)
OPA846
12
SBOS250C
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