WIDEBAND, HIGH-SENSITIVITY
TRANSIMPEDANCE DESIGN
The example of Figure 3 gives approximately 23MHz flat
bandwidth using the 0.8pF feedback compensation. If the
total output noise is bandlimited to a frequency less than the
feedback pole frequency, a simple expression for the equiva-
lent input noise current is given as Equation 3.
The high GBP and low input voltage and current noise for the
OPA846 make it an ideal wideband transimpedance ampli-
fier. Very high transimpedance gains (> 100kΩ) benefit from
the low input noise current of a JFET-input op amp, such as
the OPA657. Unity-gain stability in the op amp is not required
for application as a transimpedance amplifier. One transim-
pedance design example is shown on the front page of this
data sheet. Designs that require high bandwidths from a
large area (high capacitance) detector with relatively low
transimpedance gain will benefit from the low input voltage
noise offered by the OPA846. This input voltage noise is
peaked up over frequency at the output by the diode source
capacitance, and can, in many cases, become the limiting
factor to input sensitivity. The key elements of the design are
the expected diode capacitance (CD) with the reverse bias
voltage (–VB) applied, the desired transimpedance gain (RF),
and the GBP of the OPA846 (1750MHz). Figure 3 shows a
design using a 50pF detector diode capacitance and a 10kΩ
transimpedance gain. With these three variables set (includ-
ing the parasitic input capacitance for the OPA846 added to
CD) the feedback capacitor (CF) value can be set to control
the frequency response. To achieve a maximally flat 2nd-
order Butterworth frequency response, set the feedback pole
as shown in Equation 1.
2
2
E 2πFC
4kT
RF
EN
RF
(
)
N
D
IEQ = IN2
+
+
+
(3)
3
Where:
IEQ = equivalent input noise current if the output noise is
bandlimited to F < 1/(2πRFCF)
IN = input current noise for the op amp inverting input
EN = input voltage noise for the op amp
CD = diode capacitance
F = bandlimiting frequency in Hz (usually a post filter prior
to further signal processing)
4kT = 1.6E – 20J at T = 290K
Evaluating this expression up to the feedback pole frequency
at 16.1MHz for the circuit of Figure 3 gives an equivalent
input noise current of 4.9pA/√Hz. This is much higher than
the 2.8pA/√Hz for just the op amp. This result is dominated
by the last term in the equivalent input noise current calcu-
lation from Equation 3. It is essential in this case to use a low-
voltage noise op amp. For example, if a slightly higher input
noise voltage, but otherwise identical op amp, was used
instead of the OPA846 amplifier in this application noise
amplifier (say 2.0nV/√Hz), the total input-referred current
1
GBP
=
(1)
2πRFCF
4πRFCD
noise would increase to 7.0pA/√Hz
.
The output DC error for the circuit of Figure 3 is minimized by
including the 10kΩ to ground on the noninverting input. This
reduces the impact at the output of input bias current errors
to the offset current times the feedback resistor. To minimize
the output noise contribution of this resistor, a 0.01µF capaci-
tor is included in parallel. Worst-case output DC error for the
circuit of Figure 3 at 25°C is:
+5V
Power-supply
decoupling not shown.
10kΩ
0.01µF
VO = ID RF
OPA846
–5V
RF
10kΩ
VOS = ±0.6mV (input offset voltage) ± 0.35µA (input offset
current) • 10kΩ = ±4.1mV
λ
CD
50pF
Worst-case output offset DC drift is over the 0°C to 70°C span
is dVOS/dT = ±1.5µV/°C (input offset drift) ± 2nA/C (input
offset current drift) • 10kΩ = ±21.5µV/°C
CF
0.8pF
ID
–VB
Improved output DC precision and drift is possible, particu-
larly at higher transimpedance gains, using the JFET input of
the OPA657. The JFET input removes the input bias current
from the error equation (eliminating the need for the resistor
to ground on the noninverting input), leaving only the input
offset voltage and drift as an output error term.
FIGURE 3. Wideband, Low Noise, Transimpedance Amplifier.
Adding the common-mode and differential-mode input capaci-
tance (1.8 + 2.0)pF to the 50pF diode source capacitance of
Figure 3, with a 10kΩ transimpedance gain using the 1750MHz
GBP for the OPA846, requires a feedback pole set to 16.1MHz.
This requires a 1pF total feedback capacitance. Typical sur-
face-mount resistors have 0.2pF parasitic capacitance leaving
a required extrinsic 0.8pF value, as shown in Figure 3.
Equation 2 gives the approximate –3dB bandwidth, if CF is set
using Equation 1.
Included in the characteristic curves are transimpedance
frequency response curves for a fixed 10kΩ gain over vari-
ous detector diode capacitance settings. These curves, along
with the test circuit, are repeated in Figure 4. As the photo-
GBP
f−3dB
=
Hz
(
)
(2)
2πRFCD
OPA846
SBOS250C
11
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