TNY375-380
C5
330 pF
250 VAC
L2
D6
T1
EEL19
3.3 µH
UF4003
+12 V, 0.25 A
6
11
D1
FR106
N.C.
VR1
P6KE180A
R6
20 kΩ
1%
D2
D7
1N5819
L3
3.3 µH
F1
3.15 A
R1
FR106
L
L1
5 mH
100 Ω
+5.0 V, 0.5 A
+3.3 V, 0.5 A
C2
22 µF
400 V
1
C1
22 µF
400 V
C3
10 nF
1 kV
85-265
VAC
D8
SB340
L4
3.3 µH
7
N
4
3
R2
47 Ω
D3
1N4007
D4
1N4007
JP2
C6
C7
1000 µF
25 V
100 µF
25 V
C10
470 µF
10 V
8,9,10
12
D5
FR106
R4
C9
1000 µF
10 V
200 Ω
C5
220 µF
25 V
1/2 W
5
RTN
C8
470 µF
10 V
C11
C12
47 µF
220 µF
25 V
25 V
D
S
-12 V, 0.03 A
U2B
EN/UV
BP
TinySwitch-PK
U1
TNY376P
LTV817A
R3
1 Ω
R7
6.34 kΩ
1%
D9
UF4003
U2A
LTV817A
R5
1 kΩ
1/2 W
C4
10 µF
50 V
C14
100 nF
50 V
JP1
R9
3.3 kΩ
C13
10 µF
50 V
U3
L431
2%
R8
10 kΩ
1%
PI-4673-012009
Figure 14. TNY376PN, Four Output, 7.5 W, 13 W Peak Universal Input Power Supply.
Applications Examples
The input filter circuit (C1, L1 and C2) reduces conducted EMI. To
improve common mode EMI, this design makes use of E-ShieldTM
shielding techniques in the transformer, reducing common mode
displacement currents, and reducing EMI. These techniques,
combined with the frequency jitter of TNY376, give excellent EMI
performance, with this design achieving >10 dBmV of margin to
ENꢀꢀ022 Class B conducted EMI limits.
The circuit shown in Figure 14 is a low cost universal AC input,
four-output flyback power supply utilizing a TNY376. The
continuous output power is 7.ꢀ W with a peak of 13 W. The
output voltages are 3.3 V, ꢀ V, 12 V, and –12 V.
The rectified and filtered input voltage is applied to the primary
winding of T1. The other side of the transformer’s primary is
driven by the integrated MOSFET in U1. Diode Dꢀ, C3, R1, R2,
and VR1 compose the clamp circuit, limiting the leakage
inductance turn-off voltage spike on the DRAIN pin to a safe
value. The use of a combination Zener clamp and parallel RC
optimizes both EMI and energy efficiency.
For design flexibility, the value of C4 can be selected to pick one
of the three current limit options in U4. Doing so allows the
designer to select the current limit appropriate for the application.
•ꢀ Standard current limit is selected with a 0.1 mF BP/M pin
capacitor and is the normal choice for typical applications.
•ꢀ When a 1 mF BP/M pin capacitor is used, the current limit is
reduced, offering reduced RMS device currents and therefore
improved efficiency, but at the expense of maximum power
capability. This is ideal for thermally challenging designs where
dissipation must be minimized.
•ꢀ When a 10 mF BP/M pin capacitor is used, the current limit is
increased, extending the power capability for applications
requiring higher peak power or continuous power where the
thermal conditions allow.
Both the 3.3 V and ꢀ V outputs are sensed through resistors R6
and R7. The voltage across R8 is regulated to 2.ꢀ V by reference
IC U3. If the voltage across R8 begins to exceed 2.ꢀ V, then
current will flow in the LED inside the optocoupler U2, driven by
the cathode of U3. This will cause the transistor of the
optocoupler to sink current from the EN/UV pin of U1. When the
current exceeds the ENABLE pin threshold current, the next
switching cycle is inhibited. Conversely, when the voltage across
resistor R8 falls below 2.ꢀ V, and the current out of the ENABLE
pin is below the threshold, a conduction cycle is allowed to
occur. By adjusting the number of enabled cycles, regulation is
maintained. As the load reduces, the number of enabled cycles
decreases, lowering the effective switching frequency and
scaling switching losses with load. This provides almost
constant efficiency down to very light loads, ideal for meeting
energy efficiency requirements.
Further flexibility comes from the current limits between adjacent
TinySwitch-PK family members being compatible. The reduced
current limit of a given device is equal to the standard current limit
of the next smaller device, and the increased current limit is equal
to the standard current limit of the next larger device.
8
Rev. C 09/12
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