UC3842A, UC3843A, UC2842A, UC2843A
OPERATING DESCRIPTION
The UC3842A, UC3843A series are high performance,
is removed, or at the beginning of a soft−start interval
(Figures 24, 25). The Error Amp minimum feedback
resistance is limited by the amplifier’s source current
fixed frequency, current mode controllers. They are
specifically designed for Off−Line and DC−to−DC
converter applications offering the designer a cost effective
(0.5 mA) and the required output voltage (V ) to reach the
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solution with minimal external components.
representative block diagram is shown in Figure 18.
A
comparator’s 1.0 V clamp level:
3.0 (1.0 V) + 1.4 V
Rf(min)
≈
= 8800 ꢃ
0.5 mA
Oscillator
The oscillator frequency is programmed by the values
selected for the timing components R and C . Capacitor C
Current Sense Comparator and PWM Latch
T
T
T
The UC3842A, UC3843A operate as a current mode
controller, whereby output switch conduction is initiated by
the oscillator and terminated when the peak inductor current
reaches the threshold level established by the Error
Amplifier Output/Compensation (Pin 1). Thus the error
is charged from the 5.0 V reference through resistor R to
T
approximately 2.8 V and discharged to 1.2 V by an internal
current sink. During the discharge of C , the oscillator
T
generates and internal blanking pulse that holds the center
input of the NOR gate high. This causes the Output to be in
a low state, thus producing a controlled amount of output
signal controls the peak inductor current on
a
cycle−by−cyclebasis. The current Sense Comparator PWM
Latch configuration used ensures that only a single pulse
appears at the Output during any given oscillator cycle. The
inductor current is converted to a voltage by inserting the
deadtime. Figure 2 shows R versus Oscillator Frequency
T
and Figure 3, Output Deadtime versus Frequency, both for
given values of C . Note that many values of R and C will
T
T
T
give the same oscillator frequency but only one combination
will yield a specific output deadtime at a given frequency.
The oscillator thresholds are temperature compensated, and
the discharge current is trimmed and guaranteed to within
ground referenced sense resistor R in series with the source
S
of output switch Q1. This voltage is monitored by the
Current Sense Input (Pin 3) and compared a level derived
from the Error Amp Output. The peak inductor current under
normal operating conditions is controlled by the voltage at
pin 1 where:
10% at T = 25°C. These internal circuit refinements
J
minimize variations of oscillator frequency and maximum
output duty cycle. The results are shown in Figures 4 and 5.
In many noise sensitive applications it may be desirable to
frequency−lock the converter to an external system clock.
This can be accomplished by applying a clock signal to the
circuit shown in Figure 21. For reliable locking, the
free−running oscillator frequency should be set about 10%
less than the clock frequency. A method for multi unit
synchronization is shown in Figure 22. By tailoring the
clock waveform, accurate Output duty cycle clamping can
be achieved.
V(Pin 1) − 1.4 V
Ipk
=
3 RS
Abnormal operating conditions occur when the power
supply output is overloaded or if output voltage sensing is
lost. Under these conditions, the Current Sense Comparator
threshold will be internally clamped to 1.0 V. Therefore the
maximum peak switch current is:
1.0 V
RS
Ipk(max)
=
When designing a high power switching regulator it
becomes desirable to reduce the internal clamp voltage in
Error Amplifier
A fully compensated Error Amplifier with access to the
inverting input and output is provided. It features a typical
dc voltage gain of 90 dB, and a unity gain bandwidth of
1.0 MHz with 57 degrees of phase margin (Figure 8). The
noninverting input is internally biased at 2.5 V and is not
pinned out. The converter output voltage is typically divided
down and monitored by the inverting input. The maximum
input bias current is −2.0 ꢀ A which can cause an output
voltage error that is equal to the product of the input bias
current and the equivalent input divider source resistance.
The Error Amp Output (Pin 1) is provide for external loop
compensation (Figure 31). The output voltage is offset by
two diode drops (≈ 1.4 V) and divided by three before it
connects to the inverting input of the Current Sense
Comparator. This guarantees that no drive pulses appear at
order to keep the power dissipation of R to a reasonable
S
level. A simple method to adjust this voltage is shown in
Figure 23. The two external diodes are used to compensate
the internal diodes yielding a constant clamp voltage over
temperature. Erratic operation due to noise pickup can result
if there is an excessive reduction of the I
voltage.
clamp
pk(max)
A narrow spike on the leading edge of the current
waveform can usually be observed and may cause the power
supply to exhibit an instability when the output is lightly
loaded. This spike is due to the power transformer
interwinding capacitance and output rectifier recovery time.
The addition of an RC filter on the Current Sense Input with
a time constant that approximates the spike duration will
usually eliminate the instability; refer to Figure 27.
the Output (Pin 6) when Pin 1 is at its lowest state (V ).
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This occurs when the power supply is operating and the load
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