PRODUCT DATASHEET
AAT1141
SwitchRegTM
FastTransient 600mA Step-Down Converter
Since the inductance of a short PCB trace feeding the
input voltage is significantly lower than the power leads
from the bench power supply, most applications do not
exhibit this problem.
Dissipation due to the RMS current in the ceramic output
capacitor ESR is typically minimal, resulting in less than
a few degrees rise in hot-spot temperature.
Adjustable Output Resistor Selection
In applications where the input power source lead induc-
tance cannot be reduced to a level that does not affect
the converter performance, a high ESR tantalum or alu-
minum electrolytic should be placed in parallel with the
low ESR, ESL bypass ceramic. This dampens the high Q
network and stabilizes the system.
For applications requiring an adjustable output voltage,
the 0.6V version can be externally programmed. Resistors
R1 and R2 of Figure 5 program the output to regulate at
a voltage higher than 0.6V. To limit the bias current
required for the external feedback resistor string while
maintaining good noise immunity, the minimum sug-
gested value for R2 is 59kΩ. Although a larger value will
further reduce quiescent current, it will also increase the
impedance of the feedback node, making it more sensi-
tive to external noise and interference. Table 2 summa-
rizes the resistor values for various output voltages with
R2 set to either 59kΩ for good noise immunity or 316kΩ
for reduced no load input current.
Output Capacitor
The output capacitor limits the output ripple and pro-
vides holdup during large load transitions. A 4.7μF to
10μF X5R or X7R ceramic capacitor typically provides
sufficient bulk capacitance to stabilize the output during
large load transitions and has the ESR and ESL charac-
teristics necessary for low output ripple.
The output voltage droop due to a load transient is
dominated by the capacitance of the ceramic output
capacitor. During a step increase in load current, the
ceramic output capacitor alone supplies the load current
until the loop responds. Within two or three switching
cycles, the loop responds and the inductor current
increases to match the load current demand. The rela-
tionship of the output voltage droop during the three
switching cycles to the output capacitance can be esti-
mated by:
V
V
1.5V
0.6V
⎛
⎝
⎞
⎛
⎝
⎞
⎠
OUT
R1 =
-1 · R2 =
- 1 · 59kΩ = 88.5kΩ
⎠
REF
The adjustable version of the AAT1141, combined with
an external feedforward capacitor (C3 in Figure 1),
delivers enhanced transient response for extreme pulsed
load applications. The addition of the feedforward capac-
itor typically requires a larger output capacitor C1 for
stability.
Low Input
Current
(Without Load)
3 · ΔILOAD
=
High Noise
Immunity
COUT
V
DROOP · FS
R2 = 59kΩ
R1 (kΩ)
R2 = 316kΩ
R1 (kΩ)
VOUT (V)
Once the average inductor current increases to the DC
load level, the output voltage recovers. The above equa-
tion establishes a limit on the minimum value for the
output capacitor with respect to load transients.
0.8
0.9
1.0
1.1
1.2
1.3
1.4
1.5
1.8
2.0
2.5
3.0
3.3
19.6
29.4
39.2
49.9
59.0
68.1
78.7
88.7
88.7
137
105
158
210
267
316
365
422
475
634
732
1000
1270
1430
The internal voltage loop compensation also limits the
minimum output capacitor value to 4.7μF. This is due to
its effect on the loop crossover frequency (bandwidth),
phase margin, and gain margin. Increased output capac-
itance will reduce the crossover frequency with greater
phase margin.
187
237
267
The maximum output capacitor RMS ripple current is
given by:
Table 2: Adjustable Resistor Values For Use With
0.6V Step-Down Converter.
1
V
OUT · (VIN(MAX) - VOUT
)
IRMS(MAX)
=
·
L · F · VIN(MAX)
2 · 3
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1141.2008.07.1.4
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