AD8362
2
Moderately low resistance values should be used to minimize
scaling errors due to the 70 kΩ input resistance at the VSET
pin. This resistor string also loads the output, and it eventually
reduces the load-driving capabilities if very low values are used.
To calculate the resistor values, use
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
1
ERROR (dB –40°C)
R1 = R2' (SD/50 − 1)
(15)
0
where:
SD is the desired slope, expressed in mV/dB.
R2' is the value of R2 in parallel with 70 kΩ.
ERROR (dB +25°C)
ERROR (dB +85°C)
–1
–2
V
V
V
(+25°C)
(–40°C)
(+85°C)
OUT
OUT
OUT
For example, using R1 = 1.65 kΩ and R2 = 1.69 kΩ (R2' =
1.649 kΩ), the nominal slope is increased to 100 mV/dB.
Note, however, that doubling the slope in this manner reduces
the maximum input signal to approximately −10 dBm because
of the limited swing of VOUT (4.9 V with a 5 V power supply).
–60
–50
–40
–30
–20
–10
0
10
PIN (dBm)
Figure 56. Transfer Function and Linearity with Combined Ripple Reduction
and Temperature Compensation Circuits, Frequency = 2.14 GHz,
Single-Carrier W-CDMA, Test Model 1-64
TEMPERATURE COMPENSATION AND REDUCTION
OF TRANSFER FUNCTION RIPPLE
Because of the reduced filter capacitor, the rms voltage appearing
at the output of the error amplifier now contains significant
peak-to-peak noise. While it is critical to feed this signal back
to the VGA gain control input with the noise intact, the rms
voltage going to the external measurement node can be filtered
using a simple filter to yield a largely noise-free rms voltage.
The transfer function ripple and intercept drift of the AD8362
can be reduced using two techniques detailed in Figure 57.
CLPF is reduced from its nominal value. For broadband-
modulated input signals, this results in increased noise at
the output that is fed back to the VSET pin.
The noise contained in this signal causes the gain of the VGA
to fluctuate around a central point, moving the wiper of the
Gaussian Interpolator back and forth on the R-2R ladder.
The circuit shown in Figure 57 also incorporates a temperature
sensor that compensates temperature drift of the intercept.
Because the temperature drift varies with frequency, the amount
of compensation required must also be varied using R1 and R2.
Because the gain-control voltage is constantly moving across
at least one of taps of the Gaussian Interpolator, the relationship
between the rms signal strength of the VGA output and the
VGA control voltage becomes independent of the VGA gain
control ripple (see Figure 56). The signal being applied to the
squaring cell is now lightly AM modulated. However, this does
not change the peak-to-average ratio of the signal.
5V
These compensation techniques are discussed in more detail in
Application Note AN-653: Improving Temperature, Stability, and
Linearity of High Dynamic Range RMS RF Power Detectors.
5V
1nF
0.1µF
0.1µF
1
1kΩ
VPOS
3
2
VOUT
7
6
AD8031
V
OUT_COMP
VSET
VREF
5
AD83621
R1
R2
4
1µF
VTGT
CLPF
FREQUENCY (MHz)
900
1900
2200
R1 (kΩ)
1.02
1
R2 (kΩ)
25.5
82.5
440pF
5V
COMM
ACOM
1
19.1
0.1µF
2
1
ADDITIONAL PINS
OMITTED FOR CLARITY.
1
TMP36F
5
V
TEMP
Figure 57. Temperature Compensation and Reduction of Transfer Function Ripple
Rev. D | Page 23 of 32
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