UC3842A, 43A UC2842A, 43A
DESIGN CONSIDERATIONS
Do not attempt to construct the converter on
each succeeding cycle, alternately increasing and
decreasing the inductor current at switch turn–on. Several
oscillator cycles may be required before the inductor current
reaches zero causing the process to commence again. If
wire–wrap or plug–in prototype boards. High Frequency
circuit layout techniques are imperative to prevent pulsewidth
jitter. This is usually caused by excessive noise pick–up
imposed on the Current Sense or Voltage Feedback inputs.
Noise immunity can be improved by lowering circuit
impedances at these points. The printed circuit layout should
contain a ground plane with low–current signal and
high–current switch and output grounds returning on
separate paths back to the input filter capacitor. Ceramic
m /m is greater than 1, the converter will be unstable. Figure
2
1
19B shows that by adding an artificial ramp that is
synchronized with the PWM clock to the control voltage, the
∆I pertubation will decrease to zero on succeeding cycles.
This compensation ramp (m ) must have a slope equal to or
3
slightly greater than m /2 for stability. With m /2 slope
2
2
bypass capacitors (0.1 µF) connected directly to V , V ,
compensation, the average inductor current follows the
control voltage yielding true current mode operation. The
compensating ramp can be derived from the oscillator and
added to either the Voltage Feedback or Current Sense
inputs (Figure 32).
CC
C
and V may be required depending upon circuit layout. This
ref
provides a low impedance path for filtering the high frequency
noise. All high current loops should be kept as short as
possible using heavy copper runs to minimize radiated EMI.
The Error Amp compensation circuitry and the converter
output voltage divider should be located close to the IC and
as far as possible from the power switch and other noise
generating components.
Figure 19. Continuous Current Waveforms
(A)
∆I
Current mode converters can exhibit subharmonic
oscillations when operating at a duty cycle greater than 50%
with continuous inductor current. This instability is
independent of the regulators closed–loop characteristics
and is caused by the simultaneous operating conditions of
fixed frequency and peak current detecting. Figure 19A
Control Voltage
m1
Inductor
Current
m2
m
2
∆I + ∆I
m
m
m
2
1
2
1
m
∆
I + ∆I
m
1
Oscillator Period
shows the phenomenon graphically. At t , switch conduction
begins, causing the inductor current to rise at a slope of m .
1
This slope is a function of the input voltage divided by the
t
t
t
2
0
t
3
1
0
(B)
inductance. At t , the Current Sense Input reaches the
1
Control Voltage
m3
threshold established by the control voltage. This causes the
switch to turn off and the current to decay at a slope of m until
2
∆I
the next oscillator cycle. The unstable condition can be
shown if a pertubation is added to the control voltage,
resulting in a small ∆I (dashed line). With a fixed oscillator
period, the current decay time is reduced, and the minimum
m1
m2
Inductor
Current
Oscillator Period
current at switch turn–on (t ) is increased by ∆I + ∆I m2/m1.
2
t
t
t
The minimum current at the next cycle (t ) decreases to (∆I +
4
5
6
3
Im /m )(m /m ).Thispertubationismultipliedbym .m on
∆
2
1
2
1
2
1
10
MOTOROLA ANALOG IC DEVICE DATA
ONSEMI [ ONSEMI ]
