CS5231-3
Application Information: continued
∆V
OUT
= (∆I
LOAD
) (ESR)
This limitation directly affects load regulation. Capacitor
ESR must be minimized if output voltage must be main-
tained within tight tolerances. In such a case, it is often
advisable to use a parallel network of different types of
capacitors. For example, electrolytic capacitors provide
high charge storage capacity in a small size, while tantalum
capacitors have low ESR. The parallel combination will
result in a high capacity, low ESR network. It is also impor-
tant to physically locate the capacitance network close to
the load, and to connect the network to the load with wide
PC board traces to minimize the metal resistance.
The CS5231-3 has been carefully designed to be stable for
output capacitances greater than 10µF with equivalent
series resistance less than 1Ω. While careful board layout is
important, the user should have a stable system if these
constraints are met. A graph showing the region of stability
for the CS5231-3 is included in the “Typical Performance
Characteristics” section of this data sheet.
Input Capacitors and the V
IN
Thresholds
A capacitor placed on the V
IN
pin will help to improve
transient response. During a load transient, the input
capacitor serves as a charge “reservoir”, providing the
needed extra current until the external power supply can
respond. One of the consequences of providing this current
is an instantaneous voltage drop at V
IN
due to capacitor
ESR. The magnitude of the voltage change is again the
product of the current change and the capacitor ESR.
It is very important to consider the maximum current step
that can exist in the system. If the change in current is large
enough, it is possible that the instantaneous voltage drop
on V
IN
will exceed the V
IN
threshold hysteresis, and the IC
will enter a mode of operation resembling an oscillation.
As the part turns on, the output current I
OUT
will increase,
reaching current limit during initial charging. Increasing
I
OUT
results in a drop at V
IN
such that the shutdown
threshold is reached. The part will turn off, and the load
current will decrease. As I
OUT
decreases, V
IN
will rise and
the part will turn on, starting the cycle all over again. This
oscillatory operation is most likely at initial startup when
the output capacitance is not charged, and in cases where
the ramp-up of the V
IN
supply is slow. It may also occur
during the power transition when the linear regulator
turns on and the PFET turns off. a 15µs delay exists
between turn-on of the regulator and the AUXDRV pin
pulling the gate of the PFET high. This delay prevents
“chatter” during the power transitions. During this inter-
val, the linear regulator will attempt to regulate the output
voltage as 3.3V. If the output voltage is significantly below
3.3V, the IC will go into current limit while trying to raise
V
OUT
. It is a short-lived phenomenon and is mentioned
here to alert the user that the condition can exist. It is typi-
cally not a problem in applications. Careful choice of the
PFET switch with respect to R
DS(ON)
will minimize the volt-
age drop which the output must charge through to return
to a regulated state. More information is provided in the
section on choosing the PFET switch.
If required, using a few capacitors in parallel to increase
the bulk charge storage and reduce the ESR should give
better performance than using a single input capacitor.
Short, straight connections between the power supply and
V
IN
lead along with careful layout of the PC board ground
plane will reduce parasitic inductance effects. Wide V
IN
and V
OUT
traces will reduce resistive voltage drops.
Choosing the PFET Switch
The choice of the external PFET switch is based on two
main considerations. First, the PFET should have a very
low turn-on threshold. Choosing a switch transistor with
V
GS(ON)
≈
1V ensures the PFET will be fully enhanced with
only 3.3V of gate drive voltage. Second, the switch transis-
tor should be chosen to have a low R
DS(ON)
to minimize the
voltage drop due to current flow in the switch. The formu-
la for calculating the maximum allowable on-resistance is
R
DS(ON)(MAX)
=
V
AUX(MIN)
−
V
OUT(MIN)
1.5
×
I
OUT(MAX)
where V
AUX(MIN)
is the minimum value of the auxiliary
supply voltage, V
OUT(MIN)
is the minimum allowable out-
put voltage, I
OUT(MAX)
is the maximum output current and
1.5 is a “fudge factor” to account for increases in R
DS(ON)
due to temperature.
Output Voltage Sensing
It is not possible to remotely sense the output voltage of
the CS5231-3 since the feedback path to the error amplifier
is not externally available. It is important to minimize volt-
age drops due to metal resistance of high current PC board
traces. Such voltage drops can occur in both the supply
traces and the return traces.
The following board layout practices will help to minimize
output voltage errors:
• Always place the linear regulator as close to both load
and output capacitors as possible.
• Always use the widest possible traces to connect the lin-
ear regulator to the capacitor network and to the load.
• Connect the load to ground through the widest possible
traces.
• Connect the IC ground to the load ground trace at the
point where it connects to the load.
Current Limit
The CS5231-3 has internal current limit protection. Output
current is limited to a typical value of 850mA, even under
output short circuit conditions. If the load current drain
exceeds the current limit value, the output voltage will be
pulled down and will result in an out of regulation condi-
tion. The IC does not contain circuitry to report this fault.
Thermal Shutdown
The CS5231-3 has internal temperature monitoring circuit-
ry. The output is disabled if junction temperature of the IC
reaches a typical value of 180°C. Thermal hysteresis is typi-
7
CHERRY [ CHERRY SEMICONDUCTOR CORPORATION ]
