REF19x Series
REF19x
The requirement for a heat sink on Q1 depends on the maximum
input voltage and short-circuit current. With VS = 5 V and a 300
mA current limit, the worst-case dissipation of Q1 is 1.5 W, less
than the TO-220 package 2 W limit. However, if smaller TO-39
or TO-5 packaged devices, such as the 2N4033, are used, the
current limit should be reduced to keep maximum dissipation
below the package rating. This is accomplished by simply
raising R4.
1
2
3
4
8
7
6
5
NC
NC
V
IN
NC
1kΩ
5%
OUTPUT
+
1µF
TANT
NC
ON
10kΩ
OFF
NC = NO CONNECT
A tantalum output capacitor is used at C1 for its low equivalent
series resistance (ESR), and the higher value is required for
stability. Capacitor C2 provides input bypassing and can be an
ordinary electrolytic.
Figure 22. Membrane Switch-Controlled Power Supply
CURRENT-BOOSTED REFERENCES WITH
CURRENT LIMITING
While the 30 mA rated output current of the REF19x series is
higher than is typical of other reference ICs, it can be boosted to
higher levels, if desired, with the addition of a simple external
PNP transistor, as shown in Figure 23. Full-time current
limiting is used to protect the pass transistor against shorts.
Shutdown control of the booster stage is an option, and when
used, some cautions are needed. Due to the additional active
devices in the VS line to U1, a direct drive to Pin 3 does not
work as with an unbuffered REF19x device. To enable shutdown
control, the connection from U1 to U2 is broken at the X, and
Diode D1 then allows a CMOS control source, VC, to drive U1
to U3 for on/off operation. Startup from shutdown is not as
clean under heavy load as it is in basic REF19x series, and can
require several milliseconds under load. Nevertheless, it is still
effective and can fully control 150 mA loads. When shutdown
control is used, heavy capacitive loads should be minimized.
+V = 6V
TO 9V
(SEE TEXT)
S
Q1
TIP32A
(SEE TEXT)
OUTPUT TABLE
U1 (V)
R4
2Ω
V
OUT
REF192 2.5
REF193 3.0
REF196 3.3
REF194 4.5
REF195 5.0
R1
1kΩ
Q2
2N3906
C2
100µF
25V
R2
1.5kΩ
+
2
C3
0.1µF
F
U1
+V
3.3V
D1
OUT
S
NEGATIVE PRECISION REFERENCE WITHOUT
PRECISION RESISTORS
3
6
V
C
REF196
(SEE TABLE)
@ 150mA
1N4148
(SEE TEXT
ON SLEEP)
C1
10µF/25V
(TANTALUM)
4
R3
1.82kΩ
+
In many current-output CMOS DAC applications where the
output signal voltage must be the same polarity as the reference
voltage, it is often necessary to reconfigure a current-switching
DAC into a voltage-switching DAC using a 1.25 V reference, an
op amp, and a pair of resistors. Using a current-switching DAC
directly requires an additional operational amplifier at the
output to reinvert the signal. A negative voltage reference is
then desirable, because an additional operational amplifier is
not required for either reinversion (current-switching mode) or
amplification (voltage-switching mode) of the DAC output
voltage. In general, any positive voltage reference can be
converted into a negative voltage reference using an operational
amplifier and a pair of matched resistors in an inverting
configuration. The disadvantage to this approach is that the
largest single source of error in the circuit is the relative
matching of the resistors used.
R1
S
V
OUT
COMMON
F
V
S
COMMON
Figure 23. Boosted 3.3 V Referenced with Current Limiting
In this circuit, the power supply current of reference U1 flowing
through R1 to R2 develops a base drive for Q1, whose collector
provides the bulk of the output current. With a typical gain of
100 in Q1 for 100 mA to 200 mA loads, U1 is never required to
furnish more than a few mA, so this factor minimizes tempera-
ture-related drift. Short-circuit protection is provided by Q2,
which clamps the drive to Q1 at about 300 mA of load current,
with values as shown in Figure 23. With this separation of
control and power functions, dc stability is optimum, allowing
most advantageous use of premium grade REF19x devices for
U1. Of course, load management should still be exercised. A
short, heavy, low dc resistance (DCR) conductor should be used
from U1 to U6 to the VOUT Sense Point S, where the collector of
Q1 connects to the load, Point F.
The circuit illustrated in Figure 24 avoids the need for tightly
matched resistors by using an active integrator circuit. In this
circuit, the output of the voltage reference provides the input
drive for the integrator. To maintain circuit equilibrium, the
integrator adjusts its output to establish the proper relationship
between the reference’s VOUT and GND. Thus, any desired
negative output voltage can be selected by substituting for the
appropriate reference IC. The sleep feature is maintained in the
circuit with the simple addition of a PNP transistor and a 10 kΩ
resistor.
Because of the current limiting configuration, the dropout
voltage circuit is raised about 1.1 V over that of the REF19x
devices, due to the VBE of Q1 and the drop across Current Sense
Resistor R4. However, overall dropout is typically still low
enough to allow operation of a 5 V to 3.3 V regulator/reference
using the REF196 for U1 as noted, with a VS as low as 4.5 V and
a load current of 150 mA.
Rev. I | Page 20 of 28
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